A State-of-the-Art 13cm Television Transmitter, Part-1

Henk Medenblink, PE1JOK

A State-of-the-Art 13cm Television Transmitter, Part-1

VHF Communications 2/1998

1. Introduction

The growing popularity of frequency modulated Amateur Television has led to an increasing activity on the 23cm band. Unlike other parts of our interesting hobby, ATV is mainly driven by home made designs. This growth and activity is therefore primarily dependent on regular publications of 23cm ATV transmitter designs. However, the attractive 13cm frequency band isn’t explored yet by lots of amateurs. The increasing pressure of ISM activities on this band could lead to the amateur allocations being lost in the future. That’s the reason why it seems interesting to get some amateur activity on 13cm. With the publication of a sophisticated design a first step would be taken. The growing demands for modern components and low cost designs on mobile communication markets like GSM, DECT and PCS, have made it possible for radio amateurs to create their own professional designs. In this article the first part of a state-of-the-art 13cm ATV transmitter will be presented. This transmitter covers the entire 13cm frequency band in 1 MHz steps.

2. Block Diagram

The block diagram of the 13cm ATV transmitter is shown in Fig.1. It consists of a VCO-PLL unit, a 1.5 Watt power amplifier, video baseband processing and audio subcarrier unit and last but not least a microcontroller board with text inserter.

The microcontroller is responsible for controlling functions of the transmitter. It is possible, for example, to tune the audio carriers separately in 10 kHz increments between 5.5 and 8 MHz. Video polarity is controlled by a software menu and also frequency adjustment of the transmitter is performed with ‘Up’ and ‘Down’ tuning keys. A specially developed text inserter is controlled by the CPU board and is capable of generating a line of text through and inserting it into the composite video signal. The position of this text line and black background is determined by another software menu. The text lines are edited from a special menu and stored in EEPROM.

3. The VCO

The main part of the transmitter consists of a VCO, which is controlled by a modern PLL IC from National Semiconductor, the LMX2325. Because I have not seen any interesting designs of VCO’s covering the 2320-2450 MHz frequency range, I decided to develop one. This resulted in a VCO design which is capable of being tuned between 2250 and 2550 MHz within a 0-12V tuning voltage range and very good linearity. Above 12 volts the tuning curve becomes non-linear due to non-linearities of the varactor diode. The resulting tuning curve is given in Fig.2.

The schematic of the VCO-PLL unit is shown in Fig.3. The VCO is designed using the ‘negative resistance’ method. A Common Collector configuration with a certain emitter feedback capacitor will create a ‘negative resistance’ at the base terminal. A series resonant circuit formed by a transmission line T1, coupling capacitor C10 and varactor D1 will make the total equivalent resistance real and negative at one frequency. This occurs at the frequency of oscillation. The oscillation process starts due to noise in the circuit. The amplitude of the oscillation will stabilise due to compression effects of the transistor. At this point the total equivalent resistance is equal to zero (or loop gain equals 1).

The amount of saturation due to compression of the transistor is determined by the amount of negative resistance during start up. This amount of negative resistance is determined by feedback capacitor value C9 and the internal base-emitter capacitance Cp of the transistor Q1. The combination of these two are also responsible for the flatness of the negative resistance across the band and thus the flatness of the output power of the oscillator, which is within 1dB. Maximum output power is obtained at the high end of the frequency band. Most amplifiers have a decreasing gain at higher frequencies so that total cascaded gain will be flat.

Transistor Q2 has two functions, power gain and isolation. Isolation between an oscillator and a load is very important to maintain flatness. This can be explained by the phenomena ‘pulling’. Pulling is the change in oscillation frequency due to a variable load. This variable load can be created if the oscillator is loaded by a termination which is driven by a certain coax with a given length. If the characteristic impedance of the coax and the termination are different then a mismatch occurs. This results in a complex impedance which can be seen looking from the output terminal of the oscillator towards the load. Varying the length of the coax corresponds to a constant VSWR circle on the Smith Chart. So the magnitude of the reflection coefficient stays the same but the phase will vary between 0 and 360 degrees. This varying phase is responsible for the change in oscillation frequency if isolation between oscillator and load is bad. The same situation occurs when the coax length stays constant but the oscillation frequency is tuned.

If the coax length is not small compared to a quarter wave length, then a ripple in the tuning curve will occur if isolation is bad. Because we want to modulate the VCO with a broadband video signal and because we want to be able to tune the transmitter all over the entire frequency range, we do not want any non-linearities due to pulling. This can be overcome by an small amount of isolation and short cable lengths between VCO and power amplifier (keeping in mind that input VSWR’s of most PA’s are generally not good).

4. The Phase-Locked-Loop

Because the stability of a free running microwave VCO is not good enough, there may be a need for a form of frequency stabilization. This can be obtained by a Phase-Locked-Loop. A disadvantage of a controlling loop is the fact that they all have a certain loop bandwidth. In the case of a phase-locked-loop, disturbances with a frequency lower than this loop bandwidth will be equalized. Frequency disturbances above this loop bandwidth will not disappear.

If we want to modulate the VCO with a very broadband signal such as video, then care has to be taken with the selection of the lower edge of this loop bandwidth. Because video contains very low frequencies (50 Hz) this loop bandwidth must be lower than this frequency. In this design a bandwidth of 30 Hz is chosen.

A disadvantage of a low loop bandwidth is the speed of the loop. With the choice of 30Hz a compromise has been taken between lock time and lowest modulation frequency.

In this design a National Semiconductor LMX2325 IC fulfills the role of the PLL. This IC contains all the components needed to build a phase-locked-loop, a reference oscillator, reference divider, SHF dual modulus prescaler, swallow counter/divider and a low-leakage charge-pump phase detector. All of these functions come in a tiny TSSOP 20 package.

The current output of the phase detector is externally supplied with 5 volts. The loop filter, which consists of some capacitors and resistors, converts this current into a tuning voltage. This voltage will be limited to 5 volts due to the external phase detector supply voltage. Because the VCO tuning range varies to 12 volt, a voltage amplifier is needed to increase the DC loop gain. This amplifier (TLC272) also creates a low driving impedance towards the varactor input, which results in a broad modulation bandwidth. In Fig.4 a simulation result of the closed loop modulation response is shown.

The loop filter around the phase locked loop can easily be calculated with formulas given in National’s application note for the LMX2325. For convenience they are reproduced here in Fig.5.

The structure of this filter in combination with the charge pump output creates a Type II third order system. With the given values of C1, C2, R2, R3 and C3, which are rounded to closest practical values, the loop bandwidth is dimensioned at about 30 Hz. Because the loop gain is very high due to the VCO constant, response of the phase detector and the selected step size of 1 MHz, very high values of loop filter components would result. To overcome this problem, a very low reference frequency is selected by software. The reference divider is programmed for 25 kHz step size. This causes a high value of the programmable divider and thus a lower loop gain. The lower the loop gain, the more practical component values will result. The final tuning mechanism is adjusted by software to 1 MHz steps.

5. Assembly

A primary design goal during development was the implementation on standard low cost FR4 PCB material with a thickness of 1.6mm. Because the dimensions of the components are small, this resulted in a very small PCB layout and losses could thus be minimised. Fig.6 shows a photograph of an assembled PCB. The final layout is given in Fig.7. Note that a fully metalised bi-layer PCB and parts can be obtained from Spectra BV in Holland, whose telephone number and address are given at the end of this article. When soldering the LMX2325, too much solder can be simply removed with solder wick.

It is preferable to start by constructing the VCO and buffer. This simple sub circuit can be tested using a high frequency counter. By applying a variable voltage at the varactor input the VCO frequency can be adjusted. A 1 to 8 volt range should result in a frequency range of 2300-2450 MHz. The circuit can a be tuned slightly by positioning the varactor on the solder pads. This is equivalent to a change of effective parasitic series inductance and will therefore result in a change of the frequency range. In worst case situations one can consider incrementing or decrementing the coupling capacitor value C10 (e.g. 0p68 or 1pF instead of 0p82). If this works then the rest of the components can be placed. A metal shielding around the PCB is preferred.

A special board which carries a synthesized audio carrier unit and video baseband part has been developed and the VCO/PLL unit can be placed on this PCB. This board will be published in the next issue.